Synchronous rectifying circuit for flyback converter

ABSTRACT

A synchronous rectifying circuit for a flyback converter includes a synchronous rectifying element (Q 2 ) coupled to the secondary winding (N 2 ) of a transformer (T) and performing a synchronous rectifying operation according to an on/off operation of the synchronous rectifying element; an auxiliary inductance circuit (L 3 ) coupled to the secondary winding (N 2 ) of the transformer (T) and having an energy discharge time period shorter than that of the secondary winding (N 2 ); and a control element (Q 3 ) for turning the synchronous rectifying element (Q 2 ) off in response to the detection of termination of the energy discharge of the auxiliary inductance circuit (L 3 ).

The present disclosure relates to the subject matter contained in Japanese Patent Application No. 2002-094276 filed on Mar. 29, 2002, which is incorporated herein by reference in its entirety.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a synchronous rectifying circuit, which is a kind of switching regulators, for a flyback converter.

2. Description of the Related Art

Conventionally, in a switching regulator, a synchronous rectifying circuit is known as a rectifying circuit for attaining a high efficiency. On one hand, a self-excited type flyback converter, often referred to as the ringing choke converter (RCC), which is a kind of switching regulators is most cheap among the switching regulators of about 50 watt or less, and so employed widely in an AC adapter etc. The synchronous rectifying circuit has been tried to be applied to such a self-excited type flyback converter in various manners, but sufficient effects have not been obtained. The reason why it is difficult to apply the synchronous rectifying circuit to the self-excited type flyback converter is that it is difficult to detect the turn-off timing of a rectifying diode in addition to that the switching frequency changes largely depending on amount of a load to be coupled.

In particular, in the ZVS (zero volt switching) utilizing the resonance with the leakage inductance of a transformer which has been performed recently in the switching of the primary side, the slope of reduction of a voltage from the time point near the turning-off of a rectifying diode is gentle, so that it has been increasingly difficult to apply the synchronous rectifying circuit.

Also, when a synchronous rectifying transistor is turned on at the time of turning the rectifying diode off, a large reverse pulse current flows. Thus, not only efficiency is degraded but also the synchronous rectifying transistor is broken at the worst.

In the meantime, various kinds of synchronous rectifying circuits each using a current transformer has been proposed. Examples are disclosed in U.S. Pat. No. 3,066,727 and JP-A-2002-10639.

The synchronous rectifying circuit of the current transformer type is configured to control the switching operation of a synchronous rectifying transistor by a voltage generated at a secondary winding in accordance with an output current flowing through the primary winding of the current transformer.

However, the aforesaid conventional synchronous rectifying circuit has the following drawbacks. (1) Considerable amount of a consumption of electric power by the current transformer. (2) The high price of the current transformer itself. (3) Large amount of the switching loss caused by the slow switching speed to an off state due to the gate input capacitance of the synchronous rectifying transistor, and by the moderate curvature of the increase/decrease rate of a voltage generated at the secondary winding of the current transformer. (4) Incapability of a wide range load due to the inability to switch the synchronous rectifying transistor when the load is small and the voltage level itself generated at the secondary winding of the current transformer becomes low.

SUMMARY OF THE INVENTION

The invention was made in view of such a conventional circumference and an object of the invention is to provide a synchronous rectifying circuit for a flyback converter which can eliminate the drawbacks of the current transformer type, that is, a synchronous rectifying circuit for a flyback converter which is high in efficiency, cheap and can cope with a wide range load.

In order to achieve the above object, according to a first aspect of the invention, there is provided a synchronous rectifying circuit for a flyback converter, including: a transformer having a primary winding and a secondary winding; a synchronous rectifying element coupled to the secondary winding of the transformer and performs a synchronous rectifying operation by turning on and off; an auxiliary inductance circuit coupled to the secondary winding of the transformer and has shorter energy discharge time period in comparison with the secondary winding; and a control element for turning off the synchronous rectifying element when a termination of energy discharge of the auxiliary inductance circuit is detected.

According to the synchronous rectifying circuit for a flyback converter thus configured, the circuit can be configured so as to be small in the voltage loss and also low in the cost (almost {fraction (1/10)} of the cost of the synchronous rectifying circuit using the current transformer). Further, since the auxiliary inductance circuit shorter in the energy discharge time period than that of the secondary winding of the transformer is employed and the termination of the energy discharge of the auxiliary inductance circuit is detected by the control element thereby to turn the synchronous rectifying element off, the switching loss can be made small.

According to a second aspect of the invention, in arrangement of the synchronous rectifying circuit in the aforesaid first aspect, the synchronous rectifying element includes a field effect transistor; and the control element turns off the synchronous rectifying element by discharging electric charges accumulated in a gate of the field effect transistor.

In this configuration, since the termination of the energy discharge of the auxiliary inductance circuit is detected by the control element thereby to discharge the electric charges accumulated in the gate of the field effect transistor and turn the field effect transistor off, the switching loss can be made small.

According to a third aspect of the invention, in arrangement of the synchronous rectifying circuit in the aforesaid first or second aspect, the transformer has an auxiliary winding extending from the second winding, and further including an isolation diode element coupled between the auxiliary winding and the auxiliary inductance circuit.

In this configuration, at the time of turning-on of the synchronous rectifying element by the auxiliary inductance circuit, reminder of the energy is regenerated to the auxiliary winding through the isolation diode element.

According to a fourth aspect of the invention, in arrangement of the synchronous rectifying circuit in the aforesaid third aspect, may further include a control element coupled to the auxiliary winding and turns on the synchronous rectifying element.

In this configuration, the synchronous rectifying element is turned on by the control element coupled to the auxiliary winding.

According to a fifth aspect of the invention, in arrangement of the synchronous rectifying circuit in the aforesaid first or second aspect, may further include a control element coupled to the secondary winding and turns on the synchronous rectifying element.

In this configuration, the synchronous rectifying element is turned on by the control element coupled to the secondary winding.

BRIEF DESCRIPTION OF THE DRAWINGS

The above objects and advantages of the present invention will become more apparent by describing in detail preferred exemplary embodiments thereof with reference to the accompanying drawings, wherein:

FIG. 1 is a circuit diagram showing a synchronous rectifying circuit for a flyback converter according to a first embodiment of the invention;

FIGS. 2A to 2F are waveform diagrams showing the operation of the synchronous rectifying circuit according to the first embodiment of the invention;

FIG. 3 is a circuit diagram showing an example of a circuit configuration for automatically adjusting a voltage between terminals of a variable resistor according to a load condition in the first embodiment of the invention;

FIG. 4 is a circuit diagram showing a synchronous rectifying circuit according to a second embodiment of the invention;

FIG. 5 is a circuit diagram showing a synchronous rectifying circuit according to a third embodiment of the invention; and

FIG. 6 is a circuit diagram showing a synchronous rectifying circuit according to a fourth embodiment of the invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Referring now to the accompanying drawings, there are shown preferred embodiments of the invention.

First Embodiment:

FIG. 1 is a circuit diagram showing a synchronous rectifying circuit for a flyback converter according to a first embodiment of the invention. In the primary side of a transformer T, a power supply voltage V_(CC) is supplied to the one terminal of a primary winding N₁ (having an inductance L₁) and a drain of a switching transistor (N channel type MOS field effect transistor) Q₁ is coupled to the other terminal of the primary winding. On the other hand, in a secondary side of the transformer T, one terminal of a secondary winding N₂ (having an inductance L₂) is grounded, whilst one end of a coil L₃ constituting an auxiliary inductance circuit is coupled to the other terminal of the secondary winding through a parallel arrangement of a capacitor C₃ and a variable resistor R₃, and also a source of a transistor (N channel type MOS field effect transistor) Q₂ which is a synchronous rectifying element for performing the synchronous rectifying through an ON/OFF operation is coupled to the other terminal of the secondary winding. The coil L₃ is shorter in an energy discharge period than the secondary winding N₂. In this respect, irrespective of size relation between the inductance of the coil L₃ and the inductance of the secondary winding N₂, an energy discharge time period as an auxiliary inductance circuit is preferably shorter than an energy discharge time period of the secondary winding N₂. A drain of the transistor Q₂ is grounded through a smoothing electrolytic capacitor C₁ and also coupled to an output terminal 1 for supplying an output voltage V_(o). A Schottky diode D₃ serving as a rectifying auxiliary diode is coupled between the source and the drain of the transistor Q₂.

An auxiliary winding N_(s) is provided so as to be extended from the secondary winding N₂. That is, the one terminal of the auxiliary winding N_(s) (the number of the winding is equal to that of the secondary winding N₂ in this embodiment) is coupled to the other terminal of the secondary winding N₂. A Schottky diode D₄ serving as an isolation diode element is coupled between the other terminal of the auxiliary winding N_(s) and the other end of the coil L₃. The other terminal of the auxiliary winding Ns can be used for other power supply. A coupling point between the Schottky diode D₄ and the coil L₃ is coupled to the gate of the transistor Q₂ through a counter-current blocking diode D₅. The gate of the transistor Q2 is coupled to an emitter of a transistor (PNP transistor) Q₃ serving as a control element for turning off the synchronous rectifying transistor Q₂, whilst the collector of the transistor Q₃ is coupled to the source of the transistor Q₂. Further, a coupling point between the Schottky diode D₄ and the coil L₃ is coupled to the base of the transistor Q₃ through a differentiating capacitor C₂. A protection diode D₆ is coupled between the base and the emitter of the transistor Q₃. The transistor Q₃ serves to detect the termination of an energy discharge of the coil L₃ thereby to turn the transistor Q₂ off. An output terminal 1 is coupled to the gate of the transistor Q₁ through a control circuit (not shown) for controlling the switching operation of the transistor Q₁ on the primary side in accordance with a change of the output voltage V_(o).

Next, the operation of the synchronous rectifying circuit according to the embodiment configured in this manner will be explained. First, explanation will be made as to the switching of the transistor Q₂ from an off state to an on state. When the drain voltage V_(d1) of the switching transistor Q₁ raises at a time point t_(A) as shown in FIG. 2A, the source voltage V_(s) of the synchronous rectifying transistor Q₂ rises as shown in FIG. 2B. Thus, a voltage V_(L3) generated across a series circuit configured by the coil L₃ and the parallel arrangement of the capacitor C₃ and the variable resistor R₃ rises as shown in FIG. 2C. When the voltage V_(L3) rises, a part of electric charges caused by the energy discharge of the coil L₃ is applied to the gate of the transistor Q₂ through the diode D₅, whereby a voltage V_(GS) between the gate and the source of the transistor Q₂ rises quickly as shown in FIG. 2D thereby to turn the transistor Q₂ on. Although a current Is flowing into the transistor Q₂ and the Schottky diode D₃ rises at the time point t_(A) as shown in FIG. 2E, this current decreases gradually thereafter. A current I₃ flowing through the coil L₃ decreases gradually after the time point t_(A) (a energy discharge time period T₃ of the coil L₃) as shown in FIG. 2F.

Subsequently, explanation will be made as to the switching of the transistor Q₂ from an off state to an on state. When the voltage V_(L3) generated across the series circuit configured by the coil L₃ and the parallel arrangement of the capacitor C₃ and the variable resistor R₃ falls at a time point t_(B) simultaneous with the termination of the energy discharge of the coil L₃ as shown in FIG. 2C, the transistor Q₂ is turned on. Thus, the electric charges accumulated in the gate of the transistor Q₂ is discharged through the emitter and the collector of the transistor Q₃, whereby the voltage V_(GS) between the gate and the source of the transistor Q₂ falls quickly as shown in FIG. 2D, thereby to turn the transistor Q₂ off. That is, the termination of the energy discharge of the coil L₃ is detected by the transistor Q₃ and then the transistor Q₂ is turned off. Although the current I₃ flowing through the coil L₃ becomes 0 at the time point t_(B) as shown in FIG. 2F, the current I_(S) flowing into the Schottky diode D₃ becomes 0 at a time point t_(C) later than the time point t_(B) as shown in FIG. 2E. This is because the energy discharge time period T₃ of the coil L₃ is shorter than an energy discharge time period T₂ of the secondary winding N₂ of the transformer T.

Next, the reason why a relation that the energy discharge time period T₃ is shorter than the energy discharge time period T₂ (T₃<T₂) is always satisfied will be explained by using expressions. First, energy P₁ accumulated in the primary winding N₁ (inductance L₁) of the transformer T at the moment that the switching transistor Q₁ is changed from an on state to an off state is represented by the following expression (1);

P ₁=(½)(V _(cc) ² /L ₁)T ₁ ²  (1)

where T₁ represents an ON time period of the transistor Q₁. When the energy P₁ is seen from the secondary side of the transformer T, the energy P₁ is equal to energy P2 accumulated in the secondary winding N₂ (inductance L₂) and L₂=(N₂/N₁)²L₁. Thus, the energy P₁ is represented by the following expression (2).

P ₁ =P ₂=(½)(1/L ₂){(N ₂ /N ₁)V _(cc)}² T ₁ ²  (2)

Similarly, the energy P₃ accumulated in the coil L₃ (inductance L₃) is represented by the following expression (3);

P ₃=(½)(1/L ₃){(N ₅ /N ₁)V _(cc) −ΔV} ² T ₁ ²  (3)

where ΔV represents the total of the forward voltage V_(F) of the Schottky diode D₄, a voltage between both the terminals of the variable resistor R₃ and the voltage drop across the winding resistance of the coil L₃. A part of the energy P₃ is supplied as electric charges to the gate of the synchronous rectifying transistor Q₂ through the diode D₅ thereby to turn the transistor Q₂ on. The reminder of the accumulated energy is regenerated to the auxiliary winding N_(s) through the Schottky diode D₄. Thus, energy required for turning the transistor Q₂ on can be made minimum.

Next, the energy discharge time period T₂ of the secondary winding N₂ and the energy discharge time period T₃ of the coil L₃ will be obtained. The ON resistance value of the transistor Q₂ is sufficiently small. Thus, if such an ON resistance is ignored, the voltage between both the terminals of the secondary winding N₂ is equal to the output voltage Vo and the energy P₂ is equal to the discharge energy of the secondary winding N₂, and so the energy P₂ is represented by the following expression (4). $\begin{matrix} \begin{matrix} {P_{2} = {\left( {1/2} \right)\left( {1/L_{2}} \right)\left\{ {\left( {N_{2}/N_{1}} \right)V_{cc}} \right\}^{2}T_{1}^{2}}} \\ {= {\left( {1/2} \right)\left( {1/L_{2}} \right)V_{o}^{2}T_{2}^{2}}} \end{matrix} & (4) \end{matrix}$

According to this expression (4), the energy discharge time period T₂ of the secondary winding N₂ is represented by the following expression (5).

T ₂=(N ₂ /N ₁)(V _(cc) /V _(o))T ₁  (5)

Similarly, since the voltage between both the terminals of the auxiliary winding N_(s) is (N_(s)/N₂)V_(o), the energy P₃ is represented by the following expression (6); $\begin{matrix} \begin{matrix} {P_{3} = {\left( {1/2} \right)\left( {1/L_{3}} \right)\left\{ {{\left( {N_{s}/N_{1}} \right)V_{cc}} - {\Delta \quad V}} \right\}^{2}T_{1}^{2}}} \\ {= {\left( {1/2} \right)\left( {1/L_{3}} \right)\left\{ {{\left( {N_{s}/N_{2}} \right)V_{o}} + {\Delta \quad V}} \right\}^{2}T_{3}^{2}}} \end{matrix} & (6) \end{matrix}$

where (N_(s)/N₁)V_(c)−ΔV represents an effective voltage applied to the coil L₃ within the ON time period T₁ of the transistor Q₁ and (N_(s)/N₂)V_(o)+ΔV represents an effective voltage generated at the coil L₃ within the energy discharge time period T₃ of the coil L₃. Thus, the energy discharge time period T₃ Of the coil L₃ is represented by the following expression (7). $\begin{matrix} \begin{matrix} {T_{3} = {\left\lbrack {\left\{ {{\left( {N_{s}/N_{1}} \right)V_{cc}} - {\Delta \quad V}} \right\}/\left\{ {{\left( {N_{s}/N_{2}} \right)V_{o}} + {\Delta \quad V}} \right\}} \right\rbrack T_{1}}} \\ {= {\left\lbrack {\left\{ {{\left( {N_{s}/N_{1}} \right)V_{cc}} - {\left( {N_{2}/N_{s}} \right)\Delta \quad V}} \right\}/\left\{ {V_{o} + {\left( {N_{2}/N_{s}} \right)\Delta \quad V}} \right\}} \right\rbrack T_{1}}} \end{matrix} & (7) \end{matrix}$

And by eliminating T₁ using the expression (5), the energy discharge time period T₃ of the coil L₃ is represented by the following expression (8). $\begin{matrix} \begin{matrix} {T_{3} = {\left\lbrack {\left\{ {V_{o} - {\left( {N_{1}/N_{s}} \right)\Delta \quad {V\left( {V_{o}/V_{cc}} \right)}}} \right\}/\left\{ {V_{o} + {\left( {N_{2}/N_{s}} \right)\Delta \quad V}} \right\}} \right\rbrack T_{2}}} \\ {= {\left\lbrack {\left\{ {1 - {\left( {N_{1}/N_{s}} \right)\left( {\Delta \quad {V/V_{cc}}} \right)}} \right\}/\left\{ {1 + {\left( {N_{2}/N_{s}} \right)\left( {\Delta \quad {V/V_{o}}} \right)}} \right\}} \right\rbrack T_{2}}} \end{matrix} & (8) \end{matrix}$

Thus, when ΔV is set to have a certain value, a relation that the energy discharge time period T₃ is always shorter than the energy discharge time period T₂ (T₃<T₂) can be satisfied. However, when ΔV is 0, T₃ becomes equal to T₂.

When a load to be coupled to the output terminal 1 becomes large, the ON time period of the transistor Q₁ becomes longer, and so the current I₃ flowing through the coil L₃ increases. As a result, the total voltage ΔV of the forward voltage V_(F) of the Schottky diode D₄, a voltage between both the terminals of the variable resistor R₃ and the voltage drop across the winding resistance of the coil L₃ also increases and the ratio between the energy discharge time periods T₃ and T₂ also changes. However, when the variable resistor R₃ is adjusted to adjust the voltage between both the terminals thereof, the ratio can be made almost constant, so that the synchronous rectifying can be performed in correspondence with a wide range load. In this respect, since the current I₃ is sufficient in a range of about 0.1 to 0.2 ampere, a small sized coil L₃ may be used.

In this respect, a circuit configuration as shown in FIG. 3 may be employed in order to automatically adjust the voltage between both the terminals of the variable resistor R₃ in accordance with the load condition. This circuit configuration is arranged by utilizing a fact that the switching frequency increases when the load becomes smaller in a manner that a resistor R₃ is used in place of the variable resistor R₃ (FIG. 1) and a smoothing electrolytic capacitor C₃ is used in place of the capacitor C₃ (FIG. 1). In this arrangement, the collector of the transistor (PNP transistor) Q₄ is coupled to the connection point between the coil L₃ and the resistor R₃ and the emitter of the transistor Q₄ is coupled to the one terminal of the auxiliary winding N_(s). The base of the transistor Q₄ is coupled to the one terminal of the auxiliary winding N_(s) through the capacitor C₄ and also through a series circuit of the diode D₇ and a resistor R₅ and also coupled to the other terminal of the auxiliary winding N_(s) through a resistor R₄.

In such a circuit configuration, when a triangular wave generated by the resistor R₄ and the capacitor C₄ is supplied to the base of the transistor Q₄, the transistor Q₄ performs a PWM chopper operation. When the load becomes larger, the ON time period of the transistor Q₄ becomes longer, whilst when load becomes smaller, the ON time period becomes shorter. In this manner, the voltage between both the terminals of the variable resistor R₃ (the voltage between both the terminals of the capacitor C₃) can be adjusted to a desired value automatically, so that the ratio between the energy discharge time periods T₃ and T₂ can always be made almost constant. Incidentally, the temperature compensation of the transistor Q₄ can be performed by a diode D₇.

According to the synchronous rectifying circuit of this embodiment, the drawbacks caused by the current transformer having been used conventionally can be eliminated. That is, since the embodiment does not use any current transformer, the synchronous rectifying circuit of this embodiment can be configured so as to be small in the voltage loss and also low in the cost (almost {fraction (1/10)} of the cost of the synchronous rectifying circuit using the current transformer). Further, since the embodiment is arranged in a manner that the coil L3 which is shorter in the energy discharge time period than that of the secondary winding N2 of the transformer T is employed and the termination of the energy discharge of the coil L₃ is detected by the transistor Q₃ thereby to turn the transistor Q₂ off (quickly reduce the voltage VGS between the source and the gate of the transistor Q₂), the switching loss can be made small. Thus, in the converter of about 30 to 50 watt, the synchronous rectifying circuit of the embodiment can improve the efficiency by about several percent when compared with the synchronous rectifying circuit using the current transformer. Further, since the embodiment is arranged to adjust the voltage between both the terminals of the resistor R₃ coupled to the coil L₃, the ratio between the energy discharge time period T₃ of the coil L₃ and the energy discharge time period T₂ of the secondary winding N₂ can be made almost constant and so the synchronous rectifying can be performed in correspondence with a wide range load.

Second Embodiment:

FIG. 4 is a circuit diagram showing a synchronous rectifying circuit according to a second embodiment of the invention. In the synchronous rectifying circuit according to this embodiment, a resistor R₃ is employed in place of the variable resistor R₃ (FIG. 1) and the capacitor C₃ (FIG. 1) is eliminated. The emitter of a transistor Q₅ (PNP transistor) serving as a control element for turning on the synchronous rectifying transistor Q₂ is coupled to the other terminal of an auxiliary winding N_(s), and the collector of the transistor Q₅ is coupled to the gate of the transistor Q₂. The base of the transistor Q₅ is coupled to the one terminal of the auxiliary winding N_(s) through a series circuit of a differential capacitor C₅ and a resistor R6 and also coupled to the other terminal of the auxiliary winding N_(s) through a diode D₈. The remaining configuration of this embodiment is same as that of the synchronous rectifying circuit of the aforesaid first embodiment and so the explanation thereof is omitted.

In the synchronous rectifying circuit of the second embodiment configured in this manner, the transistor Q₂ is turned on in response to the supply of electric charges to the gate of the transistor Q₂ from the transistor Q₅. In contrast, the transistor Q₂ is turned off in response to the discharge of the electric charges accumulated in the gate of the transistor Q₂ due to the cooperation of a coil L₃ and a transistor Q₃. Incidentally, since the ON time period of each of the transistors Q₅ and Q₃ is short, these transistors are not turned on simultaneously. According to this synchronous rectifying circuit, the effects similar to those of the first embodiment can be obtained.

Third Embodiment:

FIG. 5 is a circuit diagram showing a synchronous rectifying circuit according to a third embodiment of the invention. In the synchronous rectifying circuit according to this embodiment, the secondary winding of the transformer T is formed only by a secondary winding N₂ (that is, by a single winding). In the secondary side of the transformer T, the one terminal of the secondary winding N₂ is coupled to the anode of a Schottky diode D₄ and also coupled to the drain of a synchronous rectifying transistor (N channel type MOS field effect transistor) Q₂. The source of the transistor Q₂ is grounded and also coupled through a smoothing electrolytic capacitor C₁ to an output terminal 1 for supplying an output voltage V_(o). A Schottky diode D₃ serving as a rectifying auxiliary diode is coupled between the source and the drain of the transistor Q₂.

On the other hand, the other terminal of the secondary winding N₂ is coupled to the cathode of the Schottky diode D₄ through a coil L₃ and also coupled to the output terminal 1. The emitter of a transistor Q₅ (PNP transistor) serving as a turning-on control element for turning the synchronous rectifying transistor Q₂ on is coupled to the other terminal of the secondary winding N₂, and the collector of the transistor Q₅ is coupled to the gate of the transistor Q₂. The base of the transistor Q₅ is coupled to the one terminal of the secondary winding N₂ through a series circuit of a differential capacitor C₅ and a resistor R6 and also coupled to the other terminal of the secondary winding N₂ through a diode D₈. The gate of the transistor Q₂ is coupled to the collector of a transistor (NPN transistor) Q₃ and the emitter of the transistor Q₃ is grounded. A coupling point a between the coil L₃ and the Schottky diode D₄ is coupled to the base of the transistor Q₃ through a current limiting resistor R₇ and the differential capacitor C₂. A protection diode D₆ is coupled between the base and the emitter of the transistor Q₃. The remaining configuration of this embodiment is same as that of the synchronous rectifying circuit of the aforesaid first embodiment and so the explanation thereof is omitted.

In the synchronous rectifying circuit of the second embodiment configured in this manner, the transistor Q₂ is turned on in response to the supply of electric charges to the gate of the transistor Q₂ from the transistor Q₅. In contrast, the transistor Q₂ is turned off in response to the discharge of the electric charges accumulated in the gate of the transistor Q₂ due to the cooperation of the coil L₃ and the transistor Q₃. That is, the transistor Q₃ detects the termination of the energy discharge of the coil L₃ (the voltage increase at the coupling point a between the coil L₃ and the Schottky diode D₄) thereby to turn the transistor Q₂ off. Incidentally, since the ON time period of each of the transistors Q₅ and Q₃ is short, these transistors are not turned on simultaneously. According to this synchronous rectifying circuit, the effects similar to those of the first embodiment can be obtained.

In the synchronous rectifying circuits of the first to third embodiments, a general diode with a small forward voltage may be used in place of the Schottky diode D₄.

Fourth Embodiment:

FIG. 6 is a circuit diagram showing a synchronous rectifying circuit according to a fourth embodiment of the invention. In the synchronous rectifying circuit according to this embodiment, the secondary winding of the transformer T is formed only by a secondary winding N₂ (that is, by a single winding) so that two output voltages V_(o1) and V_(o2) are taken therefrom. That is, this converter operates as a forward converter with respect to the output voltage V_(o2) and as a flyback converter with respect to the output voltage V_(o1). In the secondary side of a transformer T, the one terminal of the secondary winding N₂ is coupled through a Schottky diode D₉ and a coil L₅ to an output terminal 2 for supplying the output voltage V_(o2). On the other hand, the other terminal of the secondary winding N₂ is coupled to a connection point between the Schottky diode D₉ and the coil L₅ through a Schottky diode D₁₀ and also coupled to the output terminal 2 through an electrolytic capacitor C₇. The emitter of a transistor Q₅ (PNP transistor) is coupled to a connection point between the coil L₅ and the output terminal 2 and the collector of the transistor Q₅ is coupled to the gate of a transistor Q₂. The base of the transistor Q₅ is coupled to the one terminal of the secondary winding N₂ through a series circuit of a resistor R₆ and a differential capacitor C₅. Each of a resistor R₈ and an electric charge amount limiting capacitor C₆ is coupled between the base and the emitter of the transistor Q₅. The remaining configuration of this embodiment is same as that of the synchronous rectifying circuit of the aforesaid third embodiment and so the explanation thereof is omitted.

In the synchronous rectifying circuit of this embodiment configured in this manner, the transistor Q₂ is turned on in response to the supply of electric charges to the gate of the transistor Q₂ from the transistor Q₅. In contrast, the transistor Q₂ is turned off in response to the discharge of the electric charges accumulated in the gate of the transistor Q₂ due to the cooperation of a coil L₃ and a transistor Q₃. Incidentally, since the ON time period of each of the transistors Q₅ and Q₃ is short, these transistors are not turned on simultaneously. According to this synchronous rectifying circuit, the effects similar to those of the first embodiment can be obtained.

Further, according to this synchronous rectifying circuit, the utilizing efficiency of the transformer T can be improved and the transformer T can be miniaturized as compared with the case of extracting the output voltage V_(o2) from the auxiliary winding. Furthermore, according to the third embodiment, when the output voltage V_(o) is low (low as 3.3 volt or 2.5 volt, for instance), there may arise a case that the gate voltage of the transistor Q₂ is not sufficiently high and so an ON-resistance value thereof does not reduce sufficiently. According to the synchronous rectifying circuit of this embodiment, when the transistor Q₁ is an ON state, the Schottky diode D₉ is turned on thereby to output an output voltage also serving as a high driving voltage (the output voltage V_(o2)), whereby such a problem can be eliminated. Of course, when the output voltage V_(o) is used only as the driving voltage, since a current is small, both the coil L₅ and the Schottky diode D₁₀ are not required (the coil L₅ is replaced by a resistor) and the electrolytic capacitor C₇ may be one having a small capacitance. The output voltage V_(o2) may be raised to V_(o1)+(N₂/N₁)V_(cc) at the maximum. Incidentally, according to an experiment, the high efficiency of 90 percent or more was obtained depending on the condition.

As clear from the aforesaid explanation, according to the invention, it is possible to provide a synchronous rectifying circuit for a flyback converter provided to eliminate the drawbacks of the current transformer type, that is, a synchronous rectifying circuit for a flyback converter that is high in efficiency, cheap and can cope with a wide range load.

Although the present invention has been shown and described with reference to specific preferred embodiments, various changes and modifications will be apparent to those skilled in the art from the teachings herein. Such changes and modifications as are obvious are deemed to come within the spirit, scope and contemplation of the invention as defined in the appended claims. 

What is claimed is:
 1. A synchronous rectifying circuit for a flyback converter, comprising: a transformer having a primary winding and a secondary winding; a synchronous rectifying element coupled to the secondary winding of the transformer and performs a synchronous rectifying operation by turning on and off; an auxiliary inductance circuit coupled to the secondary winding of the transformer and has shorter energy discharge time period in comparison with the secondary winding; and a control element for turning off the synchronous rectifying element when a termination of energy discharge of the auxiliary inductance circuit is detected.
 2. The synchronous rectifying circuit as claimed in claim 1, wherein the synchronous rectifying element comprises a field effect transistor; and the control element turns off the synchronous rectifying element by discharging electric charges accumulated in a gate of the field effect transistor.
 3. The synchronous rectifying circuit as claimed in claim 2, wherein the transformer has an auxiliary winding extending from the secondary winding, further comprising: an isolation diode element coupled between the auxiliary winding and the auxiliary inductance circuit.
 4. The synchronous rectifying circuit as claimed in claim 3, further comprising: a control element coupled to the auxiliary winding and turns on the synchronous rectifying element.
 5. The synchronous rectifying circuit as claimed in claim 2, further comprising: a control element coupled to the secondary winding and turns on the synchronous rectifying element.
 6. The synchronous rectifying circuit as claimed in claim 1, wherein the transformer has an auxiliary winding extending from the secondary winding, further comprising: an isolation diode element coupled between the auxiliary winding and the auxiliary inductance circuit.
 7. The synchronous rectifying circuit as claimed in claim 6, further comprising: a control element coupled to the auxiliary winding and turns on the synchronous rectifying element.
 8. The synchronous rectifying circuit as claimed in claim 1, further comprising: a control element coupled to the secondary winding and turns on the synchronous rectifying element. 